? semiconductor components industries, llc, 2001 april, 2001 rev. 0 1 publication order number: cs5171demo/d cs5171demo/d demonstration note for cs5171/3 3.3 v to 5.0 v/ 400 ma boost regulator description the cs5171/3 demo board is configured as a compact, low profile and efficient boost regulator. this board allows initial evaluation of the performance of the cs5171 (260 khz) or the cs5173 (520 khz) 1.5 a boost regulator ic. the demonstration circuit converts 3.3 v to 5.0 v with a maximum load current of 400 ma. the high integration level of the cs5171 minimizes the external component count to 10. a highfrequency oscillator built into cs5171 allows the use of all surface mount components, greatly reducing the size and height of the circuit. using a ttlcompatible pulse train, one can increase and synchronize the switching frequency to almost twice the builtin frequency. this regulator can also be put into a sleep mode. the 5.0 v output is disabled and the circuit consumes minimum current. the inherent protection features of the cs5171 ensure that the power supply can survive poweron and heavy load conditions. features ? current mode control with pulsebypulse current limit ? easy external sync function ? power down mode consuming maximum 50 m a ? small board space requiring only 0.7 0.7 in. 2 ? low profile with component height less than 0.1 in. ? high energy transfer efficiency ? excellent line and load regulation ? fast transient response ? minimum output voltage ripple ? high reliability with all ceramic capacitors figure 1. cs5171 demonstration board demonstration note http://onsemi.com
cs5171demo/d http://onsemi.com 2 cs5171 boost regulator gnd 5.0 v gnd 3.3 v ss figure 2. application diagram absolute maximum ratings pin name maximum voltage maximum current 3.3 v 6.3 v 3.0 a ss 40 v 1.0 ma 5.0 v 6.3 v 400 ma electrical characteristics (t a = 25 c, v in = 3.3 v, 60 ma i out 400 ma; unless otherwise specified.) characteristic test conditions min typ max unit output voltage i out = 60 ma i out = 400 ma 4.966 4.963 v v frequency 235 260 290 khz duty cycle i out = 60 ma i out = 400 ma 34 48 % % efficiency i out = 60 ma i out = 400 ma 82 78 % % line regulation 3.0 v < v in < 4.0 v 1.5 mv/v shutdown current v c < 0.8 v, v ss = 0 v 12 50 m a startup time from v cc = 1.0 v to i out = 400 ma 5.6 ms transient response time from i out = 200 ma to i out = 400 ma, v o at steady state 85 m s sync range 280 500 khz sync pulse transition threshold rise time = 20 ns 2.5 v ss bias current ss = 0 v ss = 3.3 v 3.0 3.0 10 10 m a m a shutdown threshold 0.6 1.3 2.0 v shutdown delay ss = 5.0 v to 0 v, v c < 0.8 v 80 300 m s pin description pin name description 3.3 v input voltage pin. connect this pin to a 3.3 v supply. the maximum voltage rating on this pin is determined by the rating of the input capacitor. 5.0 v output voltage pin. the maximum current drawn from this pin is 400 ma, limited by the inductor. the maximum voltage rating on this pin is determined by the rating of the output capacitors. ss this is a multifunction pin. apply a ttl pulse train to this pin to sync the switching frequency up to almost twice the inherent frequency. pull this pin below 1.2 v to disable the output voltage and leave the board in sleep mode. gnd ground pin. there are two ground pins for input and output.
cs5171demo/d http://onsemi.com 3 figure 3. schematic v c fb nc ss v sw pgnd agnd v cc cs5171 r3 560 r2 1.6 k c1 0.01 m f c2 220 pf r1 4.99 k c4 0.1 m f pcc1762nd c7 10 m f c3216x5rj106m c5 10 m f c3216x5rj106m v in 3.3 v gnd j1 j2 j3 1 2 3 4 8 7 6 5 l1 22 m h tp4_220 d1 ma10705ct gnd c6 10 m f c3216x5rj106m +5.0 v j4 j5 operation guidelines ? the +3.3 v (j2) and gnd (j3) input terminals are located on the left side of the board. simple alligator or banana clip connections are needed to power up the demo board. ? the ss input terminal (j1) is also located on the left side of the board. similar connections are required to input this ttlcompatible logic signal. ? the +5.0 v (j4) and gnd (j5) output terminals are located on the right side of the board. connect the load between these two terminals. never short these terminals since the ic has no way to protect against a short in a boost regulator. place the probe right on the anode of d1 to examine the output voltage. ? to examine startup, lift the ss pin above shutdown threshold. this signal will turn on the ic and enable the output. the duty cycle and switching frequency are best observed at the v sw pin. figure 4. efficiency vs. current 0.01 current (a) efficiency (%) 70 80 90 0.06 0.11 0.16 0.21 0.26 0.31 0.36 0.41 0.46 theory of operation current mode control the cs5171 incorporates a current mode control scheme, in which the pwm ramp signal is derived from the power switch current. this ramp signal is compared to the output of the error amplifier to control the ontime of the power switch. the oscillator is only used here as a fixed frequency clock to ensure a constant operational frequency. at the beginning of each switching cycle, the clock signal turns on the internal power switch and the v sw pin voltage is equal to the saturation voltage of the switch. since the v sw pin voltage is applied across the inductor, its current increases linearly. the current is sensed through an emitter resistor within the ic. when the current signal reaches the error amplifier output at the v c pin, the power switch turns off and inductor current starts to fall. the catch diode d1 is forwardbiased to conduct inductor current. the v sw pin voltage is now equal to the output voltage plus the diode forward voltage. upon the arrival of the next switching cycle, the power switch turns on again to repeat the same operation. this control scheme features several advantages over the conventional voltage mode controller. first, derived directly from the inductor, the ramp signal responds immediately to variations in line voltage, reducing the response time caused by the output filter and error amplifier delay found in voltage mode control. the second benefit comes from inherent pulsebypulse current limit by merely clamping the peak switching current. finally, since current mode commands an output current rather than voltage, the filter offers only a single pole to the feedback loop. this allows both simple compensation and high bandwidth. without discrediting its apparent merits, current mode control also comes with its own peculiar problems, namely, subharmonic oscillation at duty cycles over 50%. the
cs5171demo/d http://onsemi.com 4 cs5171 solves this problem with internal slope compensation, which adds an artificial ramp to the current signal. a proper slope is selected to improve circuit stability without sacrificing the advantages of current mode control. short circuit condition when a short circuit condition occurs in a boost circuit, the ic runs at minimum duty cycle and the diode conducts for most of the switching period. since there is no way to balance the flux of the inductor the inductor current will eventually saturate, causing excessive current to be drawn from the input power supply. since control ics do not have the means to limit output current, an external current limit circuit, such as a fuse or relay, should be added to protect the ic, inductor and catch diode. start up the boost regulator experiences start up transition on either powering up the v cc pin or pulling up the ss pin. the difference is that the former generates a high initial current, which quickly charges the output capacitors to the input voltage. the start up waveform shown in figure 5 was recorded after the 3.3 v input supply was turned on. these waveforms clearly show the various phases in this power up transition. figure 5. startup waveforms when the v cc voltage is below the minimum supply voltage, the v sw pin is in high impedance. therefore, current conducts directly from the input power source to the output through the inductor and diode. once v cc reaches approximately 1.5 v, the internal power switch briefly turns on. this is a part of the cs5171's normal operation. the turnon of the power switch accounts for the initial current swing. when the v c pin voltage rises above the threshold, the internal power switch starts to switch and a voltage pulse can be seen at the v sw pin. detecting a low output voltage at the fb pin, the builtin frequency shift feature reduces the switching frequency to a fraction of its nominal value, reducing the minimum duty cycle, which is otherwise limited by the minimum on time of the switch. the peak current during this phase is clamped by the internal current limit. when the fb pin voltage rises above 0.4 v, the frequency increases to its nominal value, and the peak current begins to decrease as the output approaches the regulation voltage. the overshoot of the output voltage is prevented by the active pullon, by which the sink current of the error amplifier is increased once an overvoltage condition is detected. the overvoltage condition is defined as when the fb pin voltage is 50 mv greater than the reference voltage. sync and shutdown a ttlcompatible logic input at the s/s pin is capable of synchronizing the cs5171 up to 500 khz. as shown in figure 6, a rising edge on the ss pin voltage turns on the power switch, and also resets the oscillator. the duty cycle of the sync pulse can vary from 10% to 90% without altering the function. a logic low sustained for typically 80 m s at the ss pin shuts down the ic and reduces the supply current to about 30 m a. when the pin is not used, leave it floating. figure 6. timing diagram of sync and shutdown v sw current ramp sync design guidelines design specifications v in = 3.3 v 10% v o = 5.0 v 5.0% maximum load = 400 ma switching frequency = 260 khz design output filter the resonant frequency of the output filter should be less than f sw /50, in order to effectively filter out the switching frequency. the inductor value is selected to keep the current ripple low according to the equation in table 1. for the purposes of this demonstration board, we will select the maximum ripple current to be 200 ma. therefore, we choose an inductor that is 22 m h. allowing a 5% tolerance on the output voltage, the maximum output voltage ripple is 25 mv. using the equation for output voltage ripple and neglecting the esr term, the capacitor must be at least 10 m f. select the output capacitor to be 20 m f. the resonant frequency is equal to: f o 1 2 (l)(c) 7.0 khz
cs5171demo/d http://onsemi.com 5 table 1. formulas for calculation of electrical parameters in a boost regulator (assuming continuous conduction mode) parameter symbol formula duty cycle d v o v f v in v o v f v sat input current i in i load 1 d average inductor current i l(ave) i in inductor current ripple i l(rp) v in v sat l d f sw peak inductor current i l(pk) i load(max) 1 d max i l(rp) 2 average switch current i sw(ave) i load d 1 d peak switch current i sw(pk) i l(pk) open switch voltage v sw v o v f average diode current i d(ave) i load peak diode current i d(pk) i l(pk) diode reverse voltage v r v o v sat output ripple voltage v o(rp) i load v o v in v o 1 f sw c o v o v in esr
output capacitor rms current i c(rms) i load v o v in v in v o = output voltage v in = input voltage i load = load current c o = output capacitance esr = output capacitor's equivalent series resistance f sw = switching frequency 260 khz v sat = power switch saturation voltage (0.6 v typical) v f = diode forward voltage, 0.5 v for schottky diodes and 0.8 v for ultrafast recovery diodes knowing the inductor value, we can calculate the following parameters using the formulas in table 1: d max = 50%; d min = 37.4%; maximum i in = 800 ma; maximum i l(ave) = 800 ma; i l(rp) = 190 ma; maximum i l(pk) = 990 ma. the inductor's saturation current rating shall be higher than i l(pk) . if ripple current is small, i l(ave) is approximately equal to the rms current and shall be less than inductor's rms current rating. coiltronics' thinpac inductor is selected for its low profile and its toroidal core for low emi. the saturation current is 1.1 a and rms current rating is 1.0 a. in a boost converter, the output capacitor sees pulsed current, which causes significant output ripple on the esr of the output capacitor. ceramic capacitors have the lowest esr compared with electrolytic and tantalum capacitors, at the expense of high cost. here two 10 m f ceramic capacitors are used in parallel to double the capacitance. the esr of these capacitors is negligible, so the output ripple comes almost entirely from the charging/discharging of the output capacitance, as shown in figure 7. the output ripple is calculated using the equation in table 1. v o(rp) = 48 mv. figure 7. output voltage ripple and inductor current select diode the diode in a boost converter conducts when the power switch is turned off. the average diode current is equal to the load current. the peak reverse voltage of the diode is equal to the output voltage. schottky diodes have fast switching speed and low forward voltage. the diode selected for this design has a 1.5 a dc current rating and reverse breakdown voltage of 30 v. an ultrafast diode is recommended for high temperature application. compensate the error amplifier the goal of frequency compensation is to achieve the best transient response and dc regulation while ensuring the stability of the system. a typical compensation network, as shown in figure 8, provides a two pole, one zero frequency characteristic. this is further illustrated in the bode plot shown in figure 9. the dc gain of a transconductance error amplifier can be calculated as follows: gain dc g m r o where: g m = error amplifier transconductance; r o = error amplifier output resistance 1.0 m w .
cs5171demo/d http://onsemi.com 6 figure 8. typical compensation network cs5171 v sw gnd r1 5.0 k c2 200 p c1 0.01 m figure 9. frequency response of typical compensation network frequency (log) f p1 gain (db) dc gain f z1 f p2 a high dc gain is desirable for achieving accurate line and load regulation. the cs5171 has dc gain of about 60 db. the next step is to pick the crossover frequency of the open loop frequency response. this frequency is typically set between f sw /10 and f sw /20. we use the 10 khz as the crossover frequency in this design. the output capacitors and load create a pole whose frequency is determined by: f z1 1 2 cr l 1 2 20 12.5 637 hz at 10 khz, this pole generates 31 db gain on the power stage. in order to make the loop crossover at 10 khz, the error amplifier must provide 31 db gain. knowing this, we can then determine the value of the resistor r1 in figure 8. r1 31 db g m 5.0 k c1 and r1 create a zero to provide adequate phase margin on the crossover frequency. here we used 0.01 m f capacitor such that the zero is located at: f z1 1 2 c1r1 3.2 khz this frequency is low enough to provide over 45 of phase lead at 10 khz to ensure stability. at the same time, the time constant of c1 r1 = 51 m s is not too slow to degrade transient performance. the response of the converter to a load step can be observed in the atypical performance characteristicso section in figure 15. the low frequency pole, f p1 , is determined by the error amplifier output resistance and c1: f z1 1 2 c1r o 15 hz the second pole, f p2 , located in the high frequency range, can be placed at the output filter's esr zero or at half the switching frequency. selection of the second pole below the switching frequency will help to filter out the switching noises. here we use 200 pf such that this pole, determined by c2 and r1, is at: f z2 1 2 c2r1 160 khz select input capacitors in boost circuits, the inductor becomes part of the input filter. in continuous conduction mode, the input current waveform is triangular and does not contain a large pulsed current. this reduces the requirements imposed on the input capacitor selection. the product of this inductor ripple current and the input capacitor's esr determines the v in ripple. in this design, a 0.1 m f ceramic cap is used in parallel with a 10 m f ceramic cap, providing a total capacitance of 10.1 m f. as discussed earlier, the esr of these capacitors is negligible, so the v in ripple is almost entirely due to the charging and discharging of the capacitors, as shown in figure 10. figure 10. full load input ripple voltage waveforms
cs5171demo/d http://onsemi.com 7 typical performance characteristics figure 11. full load operating waveforms figure 12. minimum load operating waveforms figure 13. full load 500 khz operating waveforms figure 14. minimum load 500 khz operating waveforms figure 15. transient load
cs5171demo/d http://onsemi.com 8 bill of materials ref. des. vendor part number type pc/board value c1 digikey pcc1750nd ceramic cap 1 0.01 m f c2 digikey pcc221acvctnd ceramic cap 1 220 pf c4 digikey pcc1762nd ceramic cap 1 0.1 m f c5c7 tdk c3216x5rj106m ceramic cap 3 10 m f j1 digikey 5002knd test point 1 n/a j2j5 digikey v1055 test point 4 n/a l1 coiltronics tp4_220 inductor 1 22 m h r1 digikey p4.99khctnd resistor 1 4.99 k r2 digikey p1.6kgctnd resistor 1 1.6 k r3 digikey p560gctnd resistor 1 560 d1 digikey ma10705ctnd schottky diode 1 n/a u1 on semiconductor cs5171/3 controller 1 n/a on semiconductor and are trademarks of semiconductor components industries, llc (scillc). scillc reserves the right to make changes without further notice to any products herein. scillc makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does scillc assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. atypicalo parameters which may be provided in scill c data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. all operating parameters, including atypicalso must be validated for each customer application by customer's technical experts. scillc does not convey any license under its patent rights nor the rights of others. scillc products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body , or other applications intended to support or sustain life, or for any other application in which the failure of the scillc product could create a sit uation where personal injury or death may occur. should buyer purchase or use scillc products for any such unintended or unauthorized application, buyer shall indemnify and hold scillc and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthori zed use, even if such claim alleges that scillc was negligent regarding the design or manufacture of the part. scillc is an equal opportunity/affirmative action employer. publication ordering information central/south america: spanish phone : 3033087143 (monfri 8:00am to 5:00pm mst) email : onlitspanish@hibbertco.com tollfree from mexico: dial 018002882872 for access then dial 8662979322 asia/pacific : ldc for on semiconductor asia support phone : 13036752121 (tuefri 9:00am to 1:00pm, hong kong time) toll free from hong kong & singapore: 00180044223781 email : onlitasia@hibbertco.com japan : on semiconductor, japan customer focus center 4321 nishigotanda, shinagawaku, tokyo, japan 1410031 phone : 81357402700 email : r14525@onsemi.com on semiconductor website : http://onsemi.com for additional information, please contact your local sales representative. cs5171demo/d north america literature fulfillment : literature distribution center for on semiconductor p.o. box 5163, denver, colorado 80217 usa phone : 3036752175 or 8003443860 toll free usa/canada fax : 3036752176 or 8003443867 toll free usa/canada email : onlit@hibbertco.com fax response line: 3036752167 or 8003443810 toll free usa/canada n. american technical support : 8002829855 toll free usa/canada europe: ldc for on semiconductor european support german phone : (+1) 3033087140 (monfri 2:30pm to 7:00pm cet) email : onlitgerman@hibbertco.com french phone : (+1) 3033087141 (monfri 2:00pm to 7:00pm cet) email : onlitfrench@hibbertco.com english phone : (+1) 3033087142 (monfri 12:00pm to 5:00pm gmt) email : onlit@hibbertco.com european tollfree access*: 0080044223781 *available from germany, france, italy, uk, ireland
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